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1SLUA535A–May2010–RevisedJan2020SubmitDocumentationFeedbackCopyright2010–2020,TexasInstrumentsIncorporatedUnderstandingandDesigninganActiveClampCurrentModeControlledConverterUsingtheUCC2897AApplicationReportSLUA535A–May2010–RevisedJan2020UnderstandingandDesigninganActiveClampCurrentModeControlledConverterUsingtheUCC2897ATexasInstrumentsPMP-PSPowerSupplyControlProductsABSTRACTTheUCC2897ACurrentModeActiveClampPWMControlleroffersahighlyintegratedfeaturesetresultinginprecisioncontrolrequiredforanactiveclampforwardorflybackconverter.
TheUCC2897Adatasheetcontainsallthedesigndetailsnecessaryforaccuratelyprogrammingthedevice.
However,therearesignificantdesignconsiderationsandtrade-offsuniquetotheactiveclamppowerstagethatmustbedefinedpriortosettingupthecontroldevice.
Usingtheactiveclampforwardtopologyasanexample,theclamp,powerstageandcontrolloopcompensationisdetailedinthefollowingapplicationnote,whichisintendedtocomplementtheinformationpresentedintheUCC2897Adatasheet.
ThisinformationisalsoapplicabletotheUCC2891/2/3and4.
spacersoListofFigurestitlewillgotosecondpagespacersoListofFigurestitlewillgotosecondpageContents1Introduction32ActiveClampSwitchingFundamentals42.
1t0–t1:PowerTransfer42.
2t1–t2:Resonant52.
3t2–t3;ActiveClamp.
62.
4t3–t4:QauxOFFtoQmainON73DesignSpecifications84PowerStageDesign.
94.
1OutputPowerStageDesign.
94.
2PowerTransformerConsiderations.
194.
3ActiveClampCircuit214.
4PrimaryMOSFET(QMAIN)Selection254.
5InputCapacitance294.
6CurrentSensing314.
7SummaryofPowerStageLosses.
345OptocouplerVoltageFeedback356CompensatingtheFeedbackLoop377ProgrammingtheUCC2897APWMControlIC477.
1Step1.
Oscillator477.
2Step2.
SoftStart487.
3Step3.
VDDBypassRequirements487.
4Step4.
InputVoltageMonitoring507.
5Step5.
CurrentSenseFilteringandSlopeCompensation508SchematicandListofMaterials519SuggestedDesignImprovements.
549.
1OutputSyncRectifiers.
549.
2OvercurrentShutdown.
549.
3ComponentChanges5410Conclusion5411References55www.
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com2SLUA535A–May2010–RevisedJan2020SubmitDocumentationFeedbackCopyright2010–2020,TexasInstrumentsIncorporatedUnderstandingandDesigninganActiveClampCurrentModeControlledConverterUsingtheUCC2897AListofFigures1t0tot1PowerTransferInterval.
42t1tot2:ResonantInterval53t2tot3:ActiveClampResetInterval.
64t3tot4:Qaux"OFF"toQmain"ON"Interval75ActiveClampForwardConverterPowerStage.
96OutputInductorCurrentWaveform.
97UCC2897ABootstrapBiasSupply.
118ResetandClampCapacitorasaFunctionofInputVoltageUnderSteadyStateConditions229Low-SideClampandGateDriveCircuit2310ScopeShotofQauxTurn-Off/AmainTurnOn2611ActiveClampPowerStageWithParasiticElements2712SimplifiedZVSResonantCircuit2713PrimaryPowerStageCurrentWaveforms.
2914UCC2897AResistiveCurrentSensing3115CurrentSensingWithaCurrentSenseTransformer.
3216PowerStageLossEstimate3417OptocouplerFeedbackandSecondarySideCompensator3518UCC2897AControlSchematic3719UCC2897ASimplifiedControlBlockDiagram.
3820EffectoftheMagnetizingInductanceandClampCapacitoronPartoftheControltoOutputGainLoop.
.
.
.
.
3921GainoftheOutputLoadandCapacitorasaFunctionofFrequencyandRload4022GraphShowingthePlotoftheControltoOutputGainIncorporatingallthePreviousDefinedEquations.
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.
.
.
4123ControltoOutputgain4224GainandPhaseoftheOpto-CouplerandComponents4425Type2Compensator(FinalComponentDesignValuesShown)4526Type2CompensationGainandPhase.
4627CalculatedTotalOverallLoopGainandPhase.
4628UCC2897ASetUpDiagram.
4729UCC2897ADesignExampleSchematic51ListofTables1UCC2897ADesignExampleSpecifications.
82SynchronousRectifierMOSFETSpecifications.
163UCC2897ADesignExampleListofMaterials.
52www.
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Whilethereareseveralwidelyusedtechniquesforachievingtransformerreset,theactiveclampapproachisbyfarthebestintermsofsimplicityandoptimalperformance.
ZVS(zerovoltageswitching),lowerswitchvoltagestress,extendeddutycyclerangeandreducedEMI(electro-magneticinterference)combinedwithsignificantefficiencyimprovementsarejustafewofthereasonstoconsidertheactiveclampresettechnique.
Oneofthedisadvantagesassociatedwiththeactiveclampistheneedforaprecisedutyclamp.
Ifnotclampedtosomemaximumvalue,increaseddutycyclecanresultintransformersaturationoradditionalvoltagestressonthemainswitchwhichcanbecatastrophic.
Anotherdisadvantagehasbeentheneedforanadvancedcontroltechniquetosynchronizedelaytimingbetweentheactiveclampandmainswitchgatedrive.
OneofthemanyfeaturesoftheUCC2897Aistheprogrammablemaximumdutycycleclampaccuratetowithin±3percent.
Withaprogrammabledelaytimebetweenthemainswitchandclampswitch,thedisadvantageshistoricallyassociatedwithcontrollingtheactiveclamparenon-existentwhentheUCC2897Aisusedasthecontroldevice.
TheUCC2891/2/3/4familyaddsflexabilitybyofferingthecapabilitytodriveeitheraP-channelorN-channelclampswitchineitherahigh-sideorlow-sideconfiguration.
Foranypowersupplydesign,thesuccessofmeetingasetofgivendesignspecificationsstartswithacarefullydesignedpowerstage,controlloopandfinallysettingupthePWMcontroller.
Fortheactiveclampforwardtopologytherearesomeadditionalconsiderationsthatshallbediscussedwithinthecontextofthefollowingdesignexample.
WhiletheexamplepresentedhereinhighlightstheuseoftheUCC2897APWMcontrolIC,thedesignprocedureforthepowerstage,activeclamp,controlloopandPWMset-upaswellasthetheoreticaldevelopmentpertainingtoZVSareapplicabletotheUCC2891/2/3/4familyaswell.
ActiveClampSwitchingFundamentalswww.
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References[6]and[7]presenteightdistinctswitchingintervals,delvingdeeplyintotheactiveclampcurrentcommutation.
Usingalow-sideactiveclampconfigurationasanexample,acompleteswitchingcycle,t0–t4,canbesimplifiedandexplainedbyfourdistinctswitchingintervalsasdetailedinFigure1throughFigure4.
2.
1t0–t1:PowerTransferDuringthisstatepoweristransferredtothesecondaryasthemainswitch,QMAIN,isconductingand,undertherightconditions,hasjustturnedon.
TheprimarycurrentisflowingthroughthechannelresistanceofQMAINandismadeupofthetransformermagnetizingcurrentplusthereflectedsecondarycurrent.
Onthesecondaryside,theforwardsynchronousrectifier,QF,isonandcarryingthefullloadcurrent.
Inthepreviousstate,theloadcurrentwasinthesynchronousrectifier,QR,whichisturningoffasQFisbeingturnedonsotheyarebothsubjectedtosometurn-onloss.
Figure1.
t0tot1PowerTransferIntervalwww.
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2t1–t2:ResonantThisistheresonantstatethatoccurswithineachswitchingcycle.
DuringthisstateQMAINhasturnedoffunderZVSandtheprimarycurrentremainscontinuousasitfirstchargesupthedraintosourcecapacitorsofQMAINanddischargesthedraintosourcecapacitorofQAUX.
Thenitisdivertedthroughthebodydiode,DAUX,oftheclampswitch,QAUX.
BecauseofthedirectionoftheprimarycurrentflowingthroughDAUX,QAUXmustbeaP-channelMOSFET(body-diodepointingdown)forlow-sideactiveclampapplications.
Sincethesecondaryloadcurrentisfreewheeling,thereisnoreflectedprimarycurrent,sotheonlycurrentflowingthroughDAUXisthetransformermagnetizingcurrent.
Thereforethebody-diodeconductionlossofQAUXisminimalandtheconditionsaresetforQAUXtoturnonunderZVS.
ThedelaytimebetweenQMAINturn-offandQAUXturn-on,alsoknownastheresonantperiod,distinguishestheactiveclampfromothersingleendedtransformerresetmethodologies.
Onthesecondaryside,thevoltageacrossthesecondarywindinghascollapsedasitreflectstheprimarysidevoltage.
ThisshouldresultinanearzerovoltagetransitionofthecurrentintothebodydiodeofQRasthewindingreversesvoltagepolarity.
ThetotalloadcurrentwillnowbecarriedthroughthebodydiodeofQR.
ThewindingvoltagewillcontinuetoincreaseandtheincreaseofthevoltageinthenewpolaritywillnowcauseQRtoturnon.
Forhighcurrentapplicationsthebody-diodeconductionlossofDR,canbeamajorcontributortototalpowerloss,andisoftenoneofthekeyfactorslimitinghigherfrequencyoperation.
However,theconductionofDRisalsonecessaryforQRtoturnonunderZVS.
Althoughnotpossiblewithself-drivensynchronousrectification,wewouldprefertominimizetheconductiontimeofDRideallytozero,butstillallowQRtoturn-onunderZVS.
Adetaileddescriptionofthistransitionisavailableinreferences[10].
Figure2.
t1tot2:ResonantIntervalActiveClampSwitchingFundamentalswww.
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3t2–t3;ActiveClampThisistheactiveclampstatewherethetransformerprimaryisreset.
AlthoughtheschematicofFigure3showsanimmediatereversaloftheprimarycurrent,thetransitionfrompositivetonegativecurrentflowisactuallysmoothandhadreallybegunduringthepreviousstatewhenthemagnetizingcurrenthadreacheditsmaximumpositivepeakvalue.
Ontheprimaryside,QAUXisnowfullyturned-onasthedifferencebetweentheinputvoltage,VIN,andtheclampcapacitorvoltageisnowappliedacrossthetransformerprimary.
QAUXissubjecttominimalconductionlossasonlythemagnetizingcurrentisflowingthroughthechannelresistance.
Conversely,onthesecondaryside,QRiscarryingthefullloadcurrentthroughitschannelresistanceandisexperiencinghighconductionloss.
Figure3.
t2tot3:ActiveClampResetIntervalwww.
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4t3–t4:QauxOFFtoQmainONThisisnotaresonanttransition.
DuringthisstateQAUXhasturnedoffunderZVSandthemagnetizingcurrentstartstodecreasethevoltageonthedraintosourceparasiticcapacitoroftheQAUXandQMAINFETsresultinginavoltagedecreaseontheQMAINdraintosourcecapacitanceandanegativevoltageincreaseonthedraintosourcecapacitanceofQAUX.
Thischangeinvoltageresultsinadecreaseinvoltageacrosstheprimarywindingandthechangeinvoltageisreflectedtothesecondary.
ChangingthesecondaryvoltagerequirescurrenttochangethedraintosourcecapacitanceofQFandthegatecapacitancesofbothQFandQR.
ThisresultsintheprimarysidemagnetizingcurrentbeingdivertedtothesecondarytoalterthevoltageonthegatetosourcecapacitancesofthesecondaryswitchesandslowstherateofchangeofvoltageonQMAIN.
Thismeansthatthemagnetizingcurrentistransitioningthevoltageontheprimarysidedraintosourcecapacitances,thegate-to-sourceandgate-to-draincapacitancesofQFandboththesecondarysidegatecapacitances.
IntheEVMthiswouldtakelongerthanisavailabletoachievezerovoltsacrosstheprimaryofthetransformer.
Itisimportanttonotethatevenifzerovoltswereachievedacrosstheprimary,nofurthervoltagetransitionwouldbeachievedunlessthereflectedmagnetizingcurrentweregreaterthanthecurrentthroughLo.
ThecurrentthroughLoisbeingpulledoutofthebodydiodeofQRandunlessthereflectedmagnetizingcurrentexceedsthiscurrentthiswillcontinueandwillholdthevoltageacrossthesecondaryatzerovolts.
AtthepointwhereQMAINisturnedonthevoltageonthetransformerwillbesomewherebetweenthepeakpositivevoltageontheFET(approximately2timesVin)andVin.
However,whentheFETQMAINisturnedon,themagnetizingcurrentthroughtheprimarywindingwillbeflowingfromQMAINintothetransformerandwillbesmallincomparisontotheloadcurrent.
TheonlycurrentthatwillimmediatelydischargethroughtheFETQMAINisthecapacitivechargeoftheQMAINandQAUXFETsbecauseofprimarysideleakageinductance.
Atthecompletionofthissequence,thecircuitisbackinthet0state.
ThecurrentthroughQwillrapidlybuilduptothereflectedoutputcurrentasafunctionoftheleakageinductanceandVIN.
Adetailedexplanationofthistransitionispresentedinreference[11].
Figure4.
t3tot4:Qaux"OFF"toQmain"ON"IntervalDesignSpecificationswww.
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3Voutputisdesigned.
Theconvertermustoperatefromatelecominputvoltageof36VSomeofthekeyelectricaldesignspecificationsarelistedinTable1.
Mechanically,atargetoffittingthedesignwithinanindustrystandardhalf-brickhasalsobeenimposed.
Table1.
UCC2897ADesignExampleSpecificationsPARAMETERMINTYPMAXUNITVINInputvoltagerange364872VVONInputturn-onvoltage35VOFFInputturn-offvoltage34ηFullloadefficiency85%90%DDutycycle0.
6VOOutputvoltage3.
1353.
465VΔVO(R/P)Outputvoltageripple33mVppIOOutputloadcurrent030AILIMOutputcurrentlimit32FSWSwitchingfrequency225275kHzBWControlloopbandwidth510φMPhasemargin3060DegreesTAAmbienttemperature2540°Cwww.
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Figure5.
ActiveClampForwardConverterPowerStageTheactiveclampportionofthepowerstageconsistsoftheauxiliary(AUX)switch,QAUX,andtheclampcapacitor,CCL.
BecauseQAUXisreferencedtotheprimarysideground,thisisreferredtoasalow-sideclampconfiguration.
Thedetailsoftheactiveclampcomponentsarediscussedinsection4.
3.
Fora3.
3Voutputwith30Aofoutputcurrent,synchronousrectificationisusedontheoutputsidetomaintainhighefficiencyespeciallyatmaximumloadcurrent.
Foreaseofuseandsimplicity,self-drivensynchronousrectificationischosenasshownbytheforwardrectifier,QFandthereverserectifier,QR.
TheUC2897Ahasasoftturnofffeaturethatpreventsselfoscillationofthesecondaryselfdrivensynchronousrectifiers(reference[12])duringshutdownmakingthisanattractiveoption.
Thepowerstagedesignbeginswithselectingthesecondarysideoutputcomponents.
4.
1OutputPowerStageDesignThemaximumdutycycleforaforwardconverterusingathirdwindingresetschemeisnormallylimitedto50%.
RCDclampandresonantresetforwardconverterscanslightlyexceed50percent,buttheactiveclampresetcaneasilypushthemaximumdutycycleto60percentandhasevenbeenusedashighas70percentinsomelowervoltageapplications.
Forthisexamplethemaximumdutycycle,duringnormaloperation,islimitedto60percentat36Vinput.
At72Vinputthedutycycleisapproximately30percent.
Theoutputinductor,LO,canbecalculatedbyfirstassumingamaximumallowableinductorripplecurrent,ΔILO.
Figure6.
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1.
1OutputInductorAssumingapeak-to-peakinductorripplecurrentequalto15percentofthemaximumoutputcurrent,Faraday'sLaw(1)canbeappliedtosolveforLO,asgivenby(2).
Definingthefirstsetofvariablesfortheequationswewillbeusingthroughoutthisdocumentwehave:VO=3.
3VVin_min=36WFosc_min=225kHzDmax=0.
6ΔILo=15%IO=30AVin_max=72VFosc_max=275kHzDmin=0.
3Nowusingthesevariables,wecancalculatetheminimumrequiredinductanceneededasbelow:(1)(2)Roundingupresultsinlessripplecurrentthroughtheinductor,whileroundingdownallowsmoreripplecurrentandasmallerinductorvalue.
Bearinmind,thatasΔILOisallowedtoincrease,theRMSripplecurrentintotheoutputcapacitorincreases,asdoesanyswitchinglossexperiencedbytheoutputrectifiers.
Thesearethetrade-offsthatmustbelookedatwhendecidingontheoptimalvalueofLO.
Forthisdesign,offtheshelf(OTS)planarmagneticsareusedbecauseoftheirlowmechanicalprofileandrepeatabledesigncharacteristics.
ThePA0373fromPulseisa2Hplanardesignratedat30Adc,withasaturationcurrentratingof35A.
ThePA0373alsoincludesa1:4(maintoauxiliary)coupledwindingthatcanbeusedforaprimaryreferencedbootstrapbias,VBOOT.
Using(3),theactualvalueofΔILO(4)canbeback-calculatedforthechosenvalueofLOequalto2H.
(3)(4)Acurrentof5.
133APPtranslatesto17percentofthetotalloadcurrent,whichismorethanacceptableintermsofallowableinductorripplecurrent.
Using(5)themaximumRMSinductorcurrentiscalculatedas30.
04ARMS,whichisnearlyequaltothemaximumloadcurrent.
However,forhighervaluesofΔILOthiscalculationcanserveasadesignchecktoassurethattheoutputinductorisnotoperatingnearsaturation.
WhereTonandToffaredefinedby:(5)www.
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1.
2BootstrapBiasSupplyDuringthefreewheelingperiodwhenQRisconducting,thevoltageacrosstheoutputinductorissimplytheregulatedoutputvoltage.
AndsincethePA0373usesa1:4(NBOOT)coupledwinding,anexpressioncanbewrittenrelatingVOUTtoVBOOTassumingaSchottkydiodedropof0.
5V.
WewillsettheturnsratioandforwardvoltdropoftheSchottkyasbelow:N1o=4Vfd=0.
5VThensolvingequation(6)resultsintheexpectedvoltageonVBOOT.
(6)Thecoupledwindingtechnique,showninFigure7,workswellundernormalsteadystateconditions,howevernoticefrom(6)thattheactualvalueofVBOOTisdependantuponVOUT.
Duringabnormaloperationsuchasover-currentorshortcircuitcurrentconditions,VOUTisnolongerinregulationcausingtheconvertertooperateinahiccupmodeasVBOOTdropsbelowtheundervoltagelockoutthresholdofthePWMcontroller.
IfthePWMmustremainfullyfunctionalduringfaultconditionswhereVOUTdropsoutofregulation,thenaseparateregulatedbiasvoltagemustbederivedanddedicatedtomaintainingVBOOTabovetheUCC2897Aundervoltagelockoutthreshold.
ThisisdoneintheEVMbecausethecapacitanceneededtoprovideenoughenergystorageforstartup(reference[13])wascalculated(equations7,8and9)andwasunacceptablelargeforanEVM.
However,theseparateregulatorbuscancreateproblemsinshortcircuitconditionsasthepulsebypulsecurrentlimitingpreventsthedevicefromhittingtheshutdownovercurrentlimitandgoingintohiccupmodeandthevoltagetothechipwillnotdropout.
Becauseofthis,theconverterisnotdesignedtohandlelongtermovercurrentconditionsontheoutput.
Figure7.
UCC2897ABootstrapBiasSupplyFromtheUCC2897Adatasheet,theminimumstart-upvoltageis12.
5V,andthemaximumstartupcurrentis500μAandaruncurrentof3mA.
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4Vbelowthestartupvoltage.
Tothiscurrenthastobeaddedthedrivecurrentforthepowertransistorsforthetimeitwilltaketobringtheoutputvoltageuptothedesignedvoltageandprovidepowertothedevicethroughthebootstrapcircuit.
Thisisthechargeneededtoswitchoneachgatetimesthefrequency.
TheVrefcapacitorchargeisalsosuppliedfromthiscapacitorsothatchargemustbeincludedIicistheinternalcurrentrequiredbythedeviceforoperationQgMwillbedefinedasthegatechargeoftheMAINFET.
QgAwillbedefinesasthegatechargeontheAUXFETTssisthetimerequiredtogofrom"ON"tohavingtheoutputatnominalvoltage.
CVrefisthecapacitanceofthereferencecapacitorthatischargedtoamaximumof5.
15VVhysterisisisthedifferencebetweenthestartvoltageandtheshutdownvoltage.
Qreqdwillbethechargeneededforstartup.
Itshouldbenotedthattheswitchingoftheoutputdoesnothappenuntilthesoftstartcapacitorreaches2.
5Vandthepeakwillbe5.
0V,sotheswitchingoftheoutputdevicesonlyoccursoverhalftherisetimeandthatisreflectedintheequationbydividingtheswitchingfrequencyinhalf.
(7)(8)(9)TheseequationsdidnotaccountforthevariationsintheIsscurrent.
IfTssistheminimumacceptablestarttimethentheIsstochargetheCsscapacitorwouldbethemaximumof18.
5AbutthelongestTsswouldbewhentheIsscurrentisat10.
5A.
ThiswouldresultinaTssof1.
85theminimumandresultintherequiredaCBOOTcapacitanceincreasingtoapproximately500F.
Asstatedbeforethemethodchosenwasaseparateseriesregulatorandbecauseofthat,thecapacitancearoundthedevicethatwasrequiredwasonlysufficienttoprovidethedrivecurrenttotheoutputFETsandsuppressthenoisefromtheswitching.
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1.
3OutputCapacitorTheoutputcapacitorischosenbaseduponmanyapplicationspecificvariablessuchascost,size,functionalityandavailability.
Thisexampledeterminestheminimumoutputcapacitancebaseduponanallowableoutputripplevoltageequalto1percentoftheregulatedoutputvoltage,orroughly33mVpp.
Forhalfthe"ON"timeofthemainswitchandforhalfthe"OFF"timethecurrentinpositive(greaterthantheoutputDCandincreasingtohalfthepeaktopeakvalueandthendecreasingtozero.
ThereforethechargeintothecapacitoristheresultofatriangularshapedcurrentwithapeakofhalfthepeakcurrentripplecurrentandanapproximatetimeofonedividedbytwicethefrequencyHavingalreadycalculatedtheinductorripplecurrentfrom(4),theminimumoutputcapacitanceiscalculatedfrom(10and11)andis173Fasshownin(11).
(10)(11)Thecapacitancevaluegivenby(11)onlyaffectsthecapacitivecomponentoftheoutputripplevoltage,andthefinalselectedvalueisdominatedbyRESR(OUT)andtransientconsiderations.
Limitingtheoutputripplevoltageto33mVpp,thetotalRESR(OUT)oftheoutputcapacitorneedstobelessthan(12)asgivenby(13).
(12)(13)Iftransientresponseisadesignconsideration,thentheselectionofoutputcapacitancecanbederivedfromexaminingthetransientvoltageovershoot,VOS,thatcanbetoleratedduringastepchangeinoutputloadcurrent.
Byequatingtheinductiveenergywiththecapacitiveenergy,COcanbederivedasshownbelow:(14)Foraloadstepchangefromnoloadto50percentoffullloadandlimitingthetransientvoltageovershootto3percentoftheregulatedoutputvoltage,COiscalculatedtobe672Fasshownin(14).
Two330F,6.
3VPOSCAPcapacitorsareplacedinparallelwitha10Fceramiccapacitorasagoodtradeoffbetweentransientperformance,smallsizeandcost.
The6TPD330MPOSCAPfromSanyohasamaximumRESR(OUT)of10mΩandamaximumripplecurrentratingof4.
4ARMS.
From(Equation14),noticethatCOisproportionaltoLO,whichisalsodependantuponFSWandΔILO.
Asasidenote,thisisthereasonthatinterleavedpowerstagesaresopopular.
TheripplecancellationeffectreducesΔILOallowingmuchhigherfrequencyoperationwhichinturnreducesLO.
AsmallervalueofLOresultsinasmallervalueofCO,whichgreatlyreducestheLOCOtimeconstantofthepowerstageallowingforextremelyfasttransientresponse.
UnfortunatelyforactiveclampdesignsthelimitingfactorforresponsetimeisnottheoutputL/Cbuttheprimarysidemagnetizinginductanceandtheclampcapacitorresonantfrequency,timesafactorequalto(1–D)2.
Forthisreasonthoughwewillusethiscapacitance,theoutputwillhavesignificantoverandundershootduetothelowfrequencyresponseimposedbythetopology.
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1.
4SynchronousRectifiersTherearemanyconsiderationsforappropriatelychoosingMOSFETsusedinself-drivensynchronousrectifierapplications.
Inaself-drivenapplicationtheMOSFETgate-to-sourcevoltageisideallyderiveddirectlyfromthetransformersecondary.
Asaresult,thegatedrivevoltageisnotregulatedbutinsteadvariesasafunctionoftheinputvoltageandtransformerresetvoltage,dividedbythetransformerturnsratio.
Iftheinputvoltagerangeiswiderthantwotoone,selfdrivensynchronousrectificationmaynotbeanoptionandacontroldrivensolutionshouldinsteadbeconsidered.
Therefore,agoodstartingpointistoperformaroughcalculationtodeterminewhatthetransformerturnsrationeedstobeandthenbasedupontheinputvoltagerange,thevariationinsynchronousrectifiergatedrivevoltagecanbecalculated.
Bywritinganequationforthevolt-secondsbalanceacrosstheoutputinductoranequationfortheminimumsecondaryvoltage,VS(MIN),isgivenby(15).
(15)SincethevaluefortheriseandfalltimeofQMAINandthedelaytime(asshowninFigure2andFigure4)arenotyetknown,aworstcasevalueof3percentoftheminimumtotalperiodcaninitiallybeassumedandusedtosolve(15).
(16)Knowingtheminimuminputvoltage,theresultof(16)cannowbeusedtocalculatetheprimarytosecondarytransformerturnsratioasgivenin(17).
(17)Rounding(17)downtothenextlowestintegerresultsinaturnsratioof6,assuringthattheminimumsecondaryvoltageisgreaterthantheresultdeterminedby(16).
Aswasmentionedpreviously,thegate-to-sourcevoltageofthesynchronousMOSFETsisnotregulated,sothenextstepistodeterminehowmuchtheVGSofeachMOSFETvariesforaturnsratioof6overthefullinputvoltagerange.
TheVGSofQFvariesproportionallywiththeinputvoltagedivideddownbythetransformerturnsratio.
For36VForthereverseMOSFET,QR,thegate-to-sourcevoltageisderivedfromthetransformerresetvoltagedivideddownbythetransformerturnsratio.
Uniquetotheactiveclamptopologyisthefactthattheresetvoltageisnon-linear,andthisisfurtherdiscussedinSection4.
3.
For36VSelectionofappropriateMOSFETsalsodependsuponknowingtheRMScurrentandmaximumdrain-to-sourcevoltage.
FromtheschematicshowninFigure5itisapparentthattheVGSofQFisthesameastheVDSofQR,andtheVGSofQRisthesameastheVDSofQF.
ThereforehavingalreadycalculatedwhattheVGSisforeachMOSFET,theVDSisalsonowknown.
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(18)QFmustberatedtowithstandthepeakcurrent,asdefinedby(18)andtheRMScurrent,asdefinedby(19),duringthepowertransferinterval.
Thepeakincludes2ampsovercurrentwhiletheRMSdoesnot.
(19)Conversely,thefreewheelingMOSFET,QR,mustberatedtocarrythemaximumRMScurrent,asdefinedby(20),duringtheactiveclampresetinterval.
(20)PowerStageDesignwww.
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ThecalculatedparametersforeachMOSFETaresummarizedinTable2,andthenusedtospecifythenecessaryparameters(with20%marginadded).
(1)Asdeterminedbyequations(27)and(32).
Table2.
SynchronousRectifierMOSFETSpecificationsPARAMETERQFQRCALCULATEDPARAMETERSVGS6V3A25.
1ASPECIFIEDPARAMETERSVGS(MAX)15V15VVDS(MAX)15V15VID(MAX)(IRMS)30A30ARDS(ON)ExtremelyLowExtremelyLowQGAverageAverageNumberofMOSFETs(1)22Duringturn-offthesynchronousrectifiersofanactiveclampforwardconverterswitchatnearzerovoltage.
Duringturn-on,QFexperiencessomeswitchingloss,butQRturns-onunderZVSconditions.
Becauseofthehighlevelsofaveragecurrenteachdevicemustcarry,aMOSFETwithextremelylowonresistanceshouldbeselected.
However,QFmaystillexperiencesomeswitchingloss,soitisdesirablenottoblindlyselecttheabsolutelowestRDS(ON)device,butstillpaycloseattentiontothegatechargecharacteristic.
TheHAT2165devicefromRenesashasanRDS(ON)andQGof2.
5mΩand33nCspecifiedat4.
5VVGS.
TheabsolutemaximumelectricalratingsfortheHAT2165areVDS=30V,VGS=±20VandID=55A.
ThedeviceisavailableinalowprofileLFPAKpackagewhichisathermallyenhancedversionofanindustrystandardSO8package.
Thejunctiontoambientthermalimpedanceisapproximately60°C/WwhentheLFPAKismountedona40mm*40mm,1ozcopperpad.
Designingforanambientenvironment,TA,of40°C,andplacingadesignlimitonthemaximumallowablejunctiontemperatureequalto75percentoftheabsolutemaximumjunctiontemperature,themaximumpowerdissipationthatcanbetoleratedwithinasingleLFPAKcanbeestimatedby(21).
(21)AquickcalculationofthetotalpowerdissipatedshouldbedonetodeterminehowmanyparallelMOSFETsmustbeusedforQFandQR,inordertomaintainamaximumpowerdissipationof1.
25WperMOSFET.
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1.
4.
1QFPowerLossCalculationsAllofthefollowingQFcalculationsareperformedundertheworstcaseoperatingconditionsofminimumVIN,maximumDandmaximumIO.
Fortheswitchinglosscalculationof(24),therisetime,tR(QF),canbeapproximatedby(22),assumingthatthesinkresistancebetweenthetransformerwindingandthegateofQFislessthan3Ω,andatminimumVIN,VGSisequalto6V.
Fromthemanufacturer'sdatasheet,thegatecharge,QGoftheHAT2165Hisapproximately33nC.
SincethisdeviceturnsoffunderZVS,thefalltimeisneglected.
DefiningthevariablesQg=33nCRg=3ΩVgs=6VVf=1VVds_max=5VTbdQF=50nsWegetthefollowingresults:(22)(23)(24)AndsincetheQFsynchronousrectifieristurningoffatnearZVS,thereissomebody-diodeconductionlossatturn-off.
Forthepurposeoflossestimationonly,aworstcasebody-diodeconductiontimeof50nsisareasonableestimateasappliedto(25).
(25)TheconductionlossesduetoRMScurrentflowingthroughtheMOSFETchannelresistancearestraightforwardasgivenby(26).
Theworst-casechannelresistanceisdefinedinthedatasheet.
(26)TherearealsosomesmallbutadditionallossesassociatedwithcharginganddischargingtheMOSFETgatecapacitance,butmostofthislossisrecoveredtotheoutputloadwhenself-drivensynchronousrectificationisused.
Forapplicationsusingcontroldrivensynchronousrectification,thesesamelossesaredissipatedintheMOSFETdriveraslongasthedriverimpedanceismuchgreaterthantheinternalMOSFETimpedance.
Forthisexample,gatechargelossesarethereforeneglectedforthepurposeofsizingtheQFandQRMOSFETs.
ThemaximumpowerlossforasingleQF,HAT2165HMOSFETisestimatedby(27).
(27)Powerdissipationof2.
56Wwouldresultinajunctiontemperatureof193°C,foroperationina40°Cambientexceedingthe150°Climit.
ThenumberofparallelQFMOSFETsrequiredmaintainingthe112°Cjunctiontemperaturedesignlimitisgivenby(28).
(28)Using2FETswillresultinajunctiontemperatureof117°Cat40°ontheboard.
Thisisstillwellbelowthe150°Climit.
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1.
4.
2QRPowerLossCalculationsAllofthefollowingQRcalculationsareperformedundertheworstcaseoperatingconditionsofmaximumVIN,minimumDandmaximumIO.
SincetheQRsynchronousrectifieristurningonandoffunderZVSconditions,switchinglossesareneglected.
However,thereisgreaterbody-diodeconductionlossthanfortheQFcase.
Forthepurposeoflossestimationonly,aworstcasebody-diodeconductiontimeof150nsisareasonableestimateasappliedto(29).
(29)TheconductionlossesduetoRMScurrentflowingthroughtheMOSFETchannelresistancearestraightforwardasgivenby(30).
(30)ThemaximumpowerlossestimateforasingleQR,HAT2165LFPAKMOSFETisestimatedby(31).
(31)ThenumberofparallelQRMOSFETsrequiredmaintainingthe112°Cjunctiontemperaturedesignlimitisgivenby(32).
(32)Thejunctiontemperaturemaximumshouldbenomorethan128degreesor85%.
Thoughthedesignsaystobesafeweshouldusemorethan2FETs,eachFETwillbeonly14%overtheworstcaselimitof75%ofthemaximumallowabletemperatureandshouldbeOK.
Ifthiswereaproductiondesignrootsumsquareanalysiswouldbedoneandshouldshowthatthedesignissafe.
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2PowerTransformerConsiderationsForsimplicity,thePA0810OTSplanartransformerfromPulsewaschosen.
Ratedupto140Wandmeasuringlessthan10mmhigh,thePA0810isagoodchoiceformodulepowerapplicationsrequiringlow-profilepassivecomponents.
ThePA0810usestwoprimarywindingsofsixturnseach,andtwosingleturnsecondarywindings.
Asdeterminedfrom(17),aturnsratioofsixmustbemaintainedbyconnectingthetwoprimarywindingsinparallelandthetwosecondarywindingsinparallel.
Thisreducesthedcwindingresistancebyhalf,greatlyreducingtheI2Rconductionlosses.
SincethePA0810ispartofaconfigurablefamilyofplanartransformers,itsdesignandconstructionmaynotbeoptimalforallsituations.
ManyapplicationsmightdemandmorethanispossiblefromanOTStransformersolution,suchassmallersize,fewerwindings,increasedprimarytosecondaryisolationorhigherefficiency.
At250kHzthetransformerlossesaredominatedbycoreloss,occurringfromtimevaryingfluxswingthroughthetransformer'sBHcurveandconductionloss,resultingfromtheRMScurrentflowingthroughtheplanarwindings.
Thefluxswing,ΔB,isfirstdeterminedfrom(33)containingaconstantspecifictotheeffectiveareaofthePA0810coregeometry(Kxf)isfoundinthemanufacturer'sdatasheet).
(33)Theresultof(33)cannowbeappliedto(34)(alsoavailableinthemanufacturer'sdatasheet)todeterminethecoreloss.
(34)ThecopperlossesarearesultofRMScurrentsflowingthroughtheprimaryandsecondarywindings.
Theaveragecurrentthroughthesecondarywasdefinedpreviouslyby(19).
FromthedatasheetwehaveLmagontheprimaryas86.
25H.
Thisgivesusatotalmagneticcurrentchangeof(35)PowerStageDesignwww.
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Wecandeterminethepeakcurrentintheprimaryfrom(36)ThePrimarysideRMScurrentconsistsofthesumofthecurrentswhentheQMAINisonplusthecurrentwhenQAUXison.
(37)ForthermalpurposestheRMScurrentisderivedandthevalueisexpressedin(37).
Fromthemanufacturer'sdatasheet,theDCresistancesofthetransformerprimaryandsecondary(paralleledwindings)aregivenas11.
25mΩand0.
875mΩrespectively.
ThesevaluescannowbeusedalongwiththeknowntransformerRMScurrentstocalculatetheconductionlossesasgivenby(38).
(38)Themaximumtransformerpowerlosscannowbecalculatedby(39).
(39)Fromthetemperaturecurvesgiveninthemanufacturer'sdatasheet,1.
76Woftotalpowerlossresultsinapproximately40°Criseaboveambienttemperature.
Thereforethemaximumanticipatedtemperatureofthetransformerisapproximately80°C,asgivenby(40).
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3ActiveClampCircuitTheactiveclampoperatesontheprinciplethatthemagnetizingenergyinthetransformerisbalanced.
Thismeansthatunderstableconditions,theintegralofthevoltageacrosstheprimarywindingoveracycleshouldsumtozero.
WhentheQMAINisonthevoltageacrosstheprimaryisequaltothevoltageontheinput,Vin.
ThisvoltageispresentforthetimetheQMAINisonwhichcorrespondstothedutycycle"D".
Forasteadystateconditionthiscanbeexpressedas(41)WhentheQMAINturnsoff,ifthesystemisinbalancethevoltageacrosstheprimarywindingis:(42)ButthereisoneterminaloftheprimarystillconnectedtotheVinline.
ThismeansthatundersteadystateconditionsthevoltageacrosstheclampcapacitorwhentheQAUXisonwillbe:(43)TherewillbeslightvariationstothisduringacycleasthemagnetizingcurrentfirstchargestheCrcapacitorandthendischargesthecapacitorbutthisisthenormalvoltagethatboththeQMAINandtheQAUXwillexperience.
However,duringtransientconditionsthisvoltagewillchangeasthecontrollooprespondstotheneedtocorrecttheoutputvoltageandthisresultsinanimbalanceofthemagnetizingcurrent.
Thisimbalanceresultsinaresonanceinthecircuitbetweentheprimarysidemagnetizinginductanceandtheclampcapacitor.
ButasthesetwocomponentsareonlyconnectedduringthetimewhenQAUXison,theresonantfrequencyhasa(1-D)2factor.
AsDisbeingchangedbythecontrolcircuit,theeffectiveresonantfrequencyischangingdynamically.
TodeterminetheactualvoltagetheFETsundersteadystateconditions,wemustlookatthedutycycleandtheassociatedvoltagesmorecarefully.
Firstdefinethedutycycleasafunctionoftheinputvoltageknowingtheturnsratioprimarytosecondaryis6:1.
(44)Thisyieldsactualdutycyclesrangingfrom0.
275to0.
55.
Thesedonotcorrespondtothedefinedminimumandmaximumwehavebeenworkingwith.
Sincethecalculatedmaximumislessthanthedefinedmaximumweshouldhavenoproblem.
Next,wehavetocalculatethevoltagethatwillappearacrosstheprimaryofthetransformerwhenQMAINisoffasafunctionoftheinputvoltageanddutycycle.
ThenthevoltageacrosstheclampcapacitorCrcanbecalculated.
ThevoltageacrosstheclampcapacitorwillbethesumoftheinputvoltageVinandthevoltageacrosstheprimarywhentheQMAINisoff.
ThisisalsothevoltagethatwillbeacrosstheFETswhentheyareoff.
(45)(46)TheseequationsgivethevoltagethatwouldbeexpectedinnormaloperationandareshownintheFigure8asafunctionoftheinputvoltage.
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ResetandClampCapacitorasaFunctionofInputVoltageUnderSteadyStateConditionsHowever,duringtransienteventstheresonancebetweentheCrcapacitorandtheprimarysidemagnetizingcurrentwillresultinatransientvoltageappearingacrosstheCrcapacitor(reference[8])andhenceacrossthetransistors.
ToallowforthisvariationitissafertoassumethatthevoltageacrossthecapacitorandFETswillbesignificantlygreaterthanthevaluesdefinedinequation46andshowninthegraph.
Sinceweareusing150voltFETsweshouldhaveenoughmarginontheprimarysidebutwealsohavetocheckthesecondaryside.
Thesecondarysidegatevoltagelimitsare20V.
Toimpactthegatetheprimarysidevoltageacrossthewindingwouldhavetoapproach120volts.
Evenatminimuminputvoltage,thiswouldputtheprimarysideFETsintoavalanche.
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3.
1Low-SideClampGateDriveSinceithasalreadybeenestablishedthatQAUXmustbeagroundreferencedP-channeldevice,anegativegatedrivevoltageisrequiredtofullyturnthisdeviceon.
However,theUCC2897Adoesnotproduceoutputvoltagelevelsbelowgroundreference.
Usingagatedrivecircuitappliedtothelow-sideclamp,theP-channelMOSFETcanbedirectlydrivenfromtheUCC2897AasshowninFigure9.
Figure9.
Low-SideClampandGateDriveCircuitThefirsttimetheUCC2897AAUXvoltagegoespositive,theSchottkydiode,DAUX,isforwardbiasedandthecapacitor,CAUX,ischargedto–VAUXvolts.
ThecapacitorvoltagethendischargesthroughRAUX.
IfthetimeconstantofRAUXandCAUXin(47)ismuchgreaterthanthePWMperiod,thenthevoltageacrossCAUXremainsrelativelyconstantandtheresultantgatetosourcevoltageseenatQAUXis–VAUXwithapeakpositivevalueofzerovolts.
Therefore,VAUXiseffectivelyshiftedbelowgroundandisnowadequatefordrivingthegateofthegroundreferencedPchannelMOSFET,QAUX.
(47)ThevalueofCAUXisdeterminedbyarbitrarilychoosingRAUXtobe1kΩ,andsolving(48).
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3.
2SelectingTheClampCapacitorThefirstconsiderationforsizingtheclampcapacitoristoknowwhattheappropriatevoltageratingmustbeoverthefullrangeofVIN(showninFigure8).
ItisprobablywisetochooseacapacitorwithaslightlyhighervoltageratingthantheratingsoftheFETs.
Thevalueoftheclampcapacitorisprimarilychosenbasedontheamountofallowableripplevoltagethatcanbetolerated.
Also,itisassumedthatthevalueofthecapacitorislargeenoughtoapproximatetheclampvoltageasaconstantvoltagesource.
However,accordingto(46)VCrchangeswithinputvoltage.
Wheneveralinetransientorsuddenchangeindutycycleiscommanded,ittakessomefiniteamountoftimefortheclampvoltage,andthereforethetransformerresetvoltage,toadapt.
Largercapacitorvaluesresultinlessvoltageripplebutalsointroducesatransientresponselimitation.
Smallercapacitorvaluesresultinfastertransientresponse,atthecostofhighervoltagerippleandhighervoltageresonancesinstepload/lineresonances.
Ideallytheclampcapacitorshouldbeselectedtoallowsomevoltageripple,butnotsomuchastoaddadditionaldrain-to-sourcevoltagestresstoQMAINandontheterminalsoftheselfdrivensecondarysideswitches.
Anotherfactorthatmustbeconsideredisthetransientresponseofthecontrolloopasthecapacitoractinginresonancewiththemagnetizinginductanceofthetransformerprimarywillintroduceapole/zerointothecurrentfeedbackloopthatisafunctionof(1-D)2andwilllimittheresponsetimeofthefeedbackloop.
EventuallywhatyouwillbetradingoffistheswitchingFETvoltageversestheresponsetimeofthecontrolloopasthefastertheresponsethehigherthevoltageontheCrcapacitorversessaturationlimitsofthetransformer.
Allowapproximately20percentvoltageripplewhilepayingcloseattentiontoVDSofQMAINandthegatetosourcevoltagesofthesecondarysiderectifiersandthevolt-secondlimitationsofthetransformer.
AsimplifiedmethodforapproximatingCCL,istosolveforCCLsuchthattheresonanttimeconstantismuchgreaterthanthemaximumoff-time.
Whileadditionalfactorssuchasthepowerstagetimeconstantandcontrolloopbandwidthalsoaffecttransientresponse,thisapproach,statedin(49),assuresthattransientperformanceisnotcompromised,atleastfromtheactiveclampcircuitpointofview.
(49)Bysolving(49)forCCL,andmultiplyingtheresultbyafactorof10toassurethattheinequalityof(49)holdstrue,(49)canberewrittenas(50),expressingCCLintermsofknowndesignparameters:(50)OnceCCLiscalculatedby(51),thefinaldesignvaluemayvaryslightlyaftertheclampcapacitorripplevoltageismeasuredincircuit.
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4PrimaryMOSFET(QMAIN)SelectionSincetheclampvoltagehasalreadybeendeterminedfrom(46),thedrain-to-sourcevoltagestressofQMAINisalsoknown.
Figure8showsthatthemaximumvoltagestressoverthefullinputrangeshouldbelimitedto100V.
Also,thepeakdraincurrentofQMAINisknownfrom(36).
ThemaximumRMSdraincurrentoccursatminimuminputvoltageandmaximumloadcurrentandis3.
895Aasgivenby(52).
(52)ThereforeselectingaMOSFETwitha150VVDSrating(thisallowsa50%surgeforthetransientconditions)andanIDratingofatleast6.
45Ainsuresagreaterthan35percentdesignsafetymargin.
TheSi7846DPfromVishaySiliconixisa150V,6.
7A,N-channelMOSFETavailableinthermallyenhancedSO8PowerPAKpackage.
Fromthemanufacturer'sdatasheet,thetotalgatechargeisapproximately35nCandtheexpectedon-resistanceis41mΩfora12Vappliedgatedrive.
UsingtheIPRI(RMS)currentfrom(52),theconductionlossduetoprimarycurrentflowingthroughthechannelresistanceofQMAINisdeterminedfrom(53).
Notethat(37)includesthecurrentthroughQAUXaswellastheQMAINcurrent.
(53)AsexplainedinSection4.
4.
1,QMAINalwaysturnsoffunderZVS,butwillbesubjecttosometurn-onlossesfromthecapacitancesontheFETs,asrepresentedby(55).
Theinputvoltageconditionchosenwaslowinputbecauseitwasfeltthattheconductionlosseswhichareworseatlowlinewouldbeworsecase.
WewillalsobedischargingthegatetodraincapacitanceofQAUX.
Fromthechargediagramforthegateinthedatasheet,thechargeisabout30nCforachangeofVdsof75voltsand8voltsonthegate.
Thisisaneffectivecapacitanceof300pF.
(54)(55)WhenQMAINturnson,QAUXhasalreadyturnedoff.
TheprimarysidemagnetizingcurrentisrelativelylowandbecausethemagnetizingcurrentisneededtochargethedraintosourcecapacitanceofQMAIN,QAUXandthroughthetransformer,thegatesofthesecondaryFETs,thevoltageacrossQMAINdoesnotfallquickly.
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com26SLUA535A–May2010–RevisedJan2020SubmitDocumentationFeedbackCopyright2010–2020,TexasInstrumentsIncorporatedUnderstandingandDesigninganActiveClampCurrentModeControlledConverterUsingtheUCC2897AThisprocessisdescribedindetailinreference[11].
Ashortsummaryisbelow:WhentheQAUXturnsoffthemagnetizingcurrentandtheleakagecurrentarebothgoingintothesource,Vin.
AsaresultcurrentisdrawnfromthedraintosourcecapacitanceofQMAINandQAUXcausingthevoltageacrossQMAINtostarttocollapse(seeFigure10).
However,thiscausesthevoltageacrosstheprimarywindingtodecreaseandthisisreflectedacrosstothesecondary.
Thiscollapseonthesecondaryrequirescurrenttodischargethecapacitancethatisonthesecondarywindings.
ThissecondarycurrentisprovidedbythemagnetizingcurrentwhichinturnrobstheprimarysideofcurrentresultinginacontinuingdecreaseinprimarycurrentfromQMAINandQAUX.
Figure10.
ScopeShotofQauxTurn-Off/AmainTurnOnThisprocesswouldtakeanextendedtimeandwouldonlyfalltozerovoltsacrosstheprimarysobecauseofdutycyclelimitationsitisassumedthatthefullvoltageisacrossQMAINandQAUXattheinstantofturnon.
ItisalsoassumedthatthecurrentthroughtheprimarywindingisreversetothatwhichwouldbeinducedbyQMAINbeingonandthatthereissomeleakageinductancesoQMAINisturningonintoatleastazerocurrentconditionasfarasQMAINisconcerned.
ForlossesthentheonlylossesthroughQMAINwillbethecapacitivedischargeofthedraintosourcecapacitanceofQMAINitselfandofQAUXassumingthatthereisnoMillerinducedturnonofQAUX.
(56)AquickcheckofthemaximumjunctiontemperatureofQMAINiscalculatedtobe89.
4°Casshownin(65).
(57)97.
4°Cislessthan75percent(113°C)oftheabsolutemaximumjunctiontemperatureof150°C.
Nevertheless,whenlayingoutthePCB,placingadditionalcopperareaunderthedraintaboftheQMAINPowerPAKalsohelpstolowerthejunctiontemperature.
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4.
1PrimaryMOSFET(QMAIN)ZVSConsiderationsTheabilitytoZVSQMAINisoneoftheprimarymotivationsforusingtheactiveclamp.
DetailingtheconditionsforZVSfirstrequiresanunderstandingofthecontributingparasiticelementsasshowninFigure11.
Figure11.
ActiveClampPowerStageWithParasiticElementsTheconditionsforZVSarethatthedrain-to-sourcevoltagemustbezeropriortoQMAINswitchingoffandthatthevoltageacrossQAUXmustbezerobeforeQAUXswitcheson.
ThisconditionisachievedwhenthevoltageatnodeVA,showninFigure12,isresonantlydrivenfromzerovoltstoVclwithinthesettimeinterval.
Therefore,forthepurposeofZVS,thecircuitofFigure11canbereducedtoasimpleresonantcircuitasshowninFigure12.
Figure12.
SimplifiedZVSResonantCircuitPowerStageDesignwww.
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AsCOSS(QMAIN)ischargedtoVA,thevoltageacrossCrthebody-diodeofQMAINisreversebiasedandthecurrentthatwaspreviouslyflowingthroughthechannelresistanceofQMAINisnowdivertedtoCOSS(QMAIN)andCOSS(QAUX).
Thiscurrentconsistsoftwoidentifiablecomponents,theloadcurrentandthemagnetizingcurrent.
ThefirsttheloadcurrentwillcontinuetoflowuntilthevoltageonthedrainofQMAINreachesthevoltageontheinput.
Atthatpoint,itwillceasetoflow.
HoweverthemagnetizingcurrenthasnootherpathandwillcontinuetoflowdrivingthevoltageonbothdraintosourcecapacitorsupuntilthevoltageonthedrainofQAUXreachesonediodedropaboveground.
AtthispointthemagnetizingcurrentthenflowsthroughCrandthediodeofQAUX.
ThevoltageonCrhasbeenincreasingbutbecauseCRissolargecomparedtotheCOSS(QMAIN)andCOSS(QAUX)thatitcouldbeignoredduringthisshortintervaloftime.
ThevoltageonCOSS(QAUX)isonediodedropbelowgroundandQAUXcanbeturnedoninaZVSstate.
Thisisdescribedindetailinreference[10].
However,thefullloadcurrentplusthemagnetizingcurrentisflowingthroughtheleakageinductanceandthistriestodrivethecurrentthroughtheprimary.
Someofthemagnetizingcurrentwillbedivertedtobiasthesecondarysidecapacitances.
ThemajorityofthemagnetizingcurrentwillcontinuetochargethedraintosourcecapacitancesoftheprimarysideFETsuntilthedraintosourcecapacitancesvoltageofQAUXreacheszeroandthemagnetizingcurrentthenstartstogothroughtheparasiticdiodeoftheFET.
AtthispointtheAUXFETcanbeturnedon.
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5InputCapacitanceTheactiveclampforwardconverterisabuckderivedpowertopologywithapulsedACinputcurrenthavingahighinputdi/dtcontent,asshowninFigure13.
Figure13.
PrimaryPowerStageCurrentWaveformsTheDCoutputcurrentis30Ampsandifweassumea50%dutycyclewhichisusuallytheworstcaseforRMScurrentfortheinputcapacitorthistranslatesintoaDCcomponentof2.
5ampsinwhenQMAINisoffand2.
5ampsduringthe"ON"timeofQMAINbecauseofthe6to1turnsratioofthetransformer.
Theoutputinductorduringthattimehasapeaktopeakcurrentrampof3.
667ampswhichisbalancedabouttheDClevelsowhentheQMAINturnson,itisabout1.
83ampsbelowtheDCandbythetimeQMAINturnsoffitis1.
83ampsabovetheDClevel.
Thistranslatestotheprimaryas0.
611ampspeaktopeakor0.
3ampsaboveandbelowthecurrentpulseduringtheONtimeofQMAIN.
Inaddition,themagnetizingcurrentduringtheONtimeofQMAINfromtheprimarysideinductanceiscalculatedtobe1.
02Amps.
Thistoowillbebalancedaboutzeroovereachhalfofthetransitionor±0.
5amps.
TheDCcurrentintothesupplyisequaltohalfoftheDCcomponentneededbecausethatisaverageDCcurrentovera50%dutycycleof5amps.
TocalculatetheRMScurrentwenowhavetoperformanintegrationofthesquareofthecurrentoveracycledividedbythetimeofthatcycleandtakethesquarerootofthatintegral.
TheseintegralsarenotperfectlybalancedastheywouldbeinrealityforcalculatingtheRMScurrenttheyarecloseenough.
TheintegralsfortheDand1-Darecalculatedseparatelyassuming50%dutycycethensummedfortheRMScurrent.
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16ARMScapacitorcurrent.
Forinitiallychoosingtheinputcapacitoritisassumedthatthechangeinripplevoltageiscapacitivedominant,althoughathigherfrequencyoperationLESLandRESR(IN)candominateoverCIN.
Theminimumrequiredinputcapacitancethatwouldlimitthevoltagerippleto5%oftheminimuminputvoltageisgivenby(62).
Itisassumedthattheworstcaseripplecurrentwilloccuratapproximately50%dutycycle.
Thechargecomingoutofthecapacitorduringthetimethemainswitchisonandassumingthattheinputcurrentremainconstantisdefinedin(61).
(61)Thischargeshouldnotresultinavoltagechangeacrossthecapacitorofmorethan5%oftheminimuminputvoltage.
(62)Becausetheamountofinputripplevoltageislargecomparedtothecapacitorripplecurrent,theRESR(IN)oftheinputcapacitorislessofaconcernthanfortheoutputcapacitor.
Nonetheless,theminimumrequiredRESR(IN)shouldstillbecheckedby(63).
(63)ForamaximumVINof72V,multilayerceramicisthemostviablecapacitorchoice.
UsingtwoormoreparallelceramiccapacitorseasilysatisfiestheRESR(IN)requirementfrom(63)whilealsointroducingminimalparasiticinductance.
TheC4532X7R2A225isa2.
2μF,100VmultilayerceramiccapacitorfromTDKratedfor2.
5ARMSat300kHz,withanRESR(IN)of4mΩ.
Threeparallelcapacitorsarechosengivingatotalinputcapacitanceof6.
6F.
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6CurrentSensingTheUCC2897Ahastwocurrentsensethresholds.
Thepulsebypulsecurrentthresholdminimumvoltageis0.
43voltsandcangoashighas0.
53volts.
Thisisthethresholdthatmustbemetatfullload.
Thehiccupmodethresholdis0.
71to0.
81volts.
Inmostcasesthiswillneverbeencounteredasthepulsebypulsecurrentthresholdturnstheswitchoffbeforeitcanbereached.
ThegoalofcurrentmodecontrolistomodulatetheONtimeofQMAINbasedupontheerrorvoltageandthecurrentflowingintheoutputinductor.
Becausetheoutputcurrentissohigh,currentsensingisdoneontheprimarysidewheretheswitchedloadcurrentisreducedbythetransformerturnsratio.
PrimarysidecurrentsensingcanbedoneusingeitherasmallcurrentsenseresistorplacedinserieswiththesourceofQMAINoracurrentsensetransformer.
Whendesigningforhighefficiency,thetotallossesassociatedwitheachapproachshouldbeconsidered.
TheresistivecurrentsensingapproachisshowninFigure14,alongwiththeapproximatevoltagewaveformseenacrossthecurrentsenseresistor.
Figure14.
UCC2897AResistiveCurrentSensingFrom(36),thepeakprimarycurrentis5.
93AforIOUT=30Aplushalfoftheoutputinductorripplecurrent,butforsettingthecurrentlimit,thepeakprimarycurrentisequalto6.
25AcorrespondingtoILIM=32Awhenyouincludetheprimarysidemagnetizingcurrent.
ThevalueofRCSisgivenby(64).
(64)UsingtheprimaryRMScurrentof4.
42Afrom(37),themaximumpowerdissipatedinthenominalcurrentsenseresistorisgivenby(65).
(65)PowerStageDesignwww.
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0Winthecurrentsenseresistorwouldresultinanoverallefficiencypenaltyofabout1%.
TheimpactofthisapproachshouldbecomparedtousingacurrentsensetransformerasshowninFigure15.
Figure15.
CurrentSensingWithaCurrentSenseTransformerConsiderthecurrentsensetransformer,TCS,showninFigure15.
ThecurrentflowingthroughRCSistheprimarycurrent,IPRI,reducedbythecurrentsensetransformerturnsratio.
Foracurrenttransformerturnsratioof100to1,ICSduringpeakcurrentlimitisdeterminedby(66).
(66)AndfromICS(CL_PK),thecurrentsensingresistoriscalculatedby(67).
(67)UsingtheprimaryRMScurrentof4.
42Afrom(37),themaximumpowerdissipatedinthe6.
9Ωcurrentsenseresistorisgivenby(68).
(68)BecauseoftheearlyreleaseoftheEVMthecurrentsenseresistorontheEVMis4.
64Ωandthecurrentlimitpointisapproaching44A.
www.
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Thelargestcontributionofpowerdissipationcomesfromtheprimarycurrentflowingthroughthesingleturndcresistance.
FortheP8208,thedcresistancesare6mΩforthesingleturnprimaryand5.
5Ωforthe100turnsecondary.
Thecurrentsensetransformerconductionlossesaregivenby(69)and(70).
(69)(70)TheSchottkyrectifierusedinthesensingcircuitofFigure15,alsoaddsasmallamountofpowerdissipationasaproductoftheRMScurrentanddiodevoltagedropwhenthediodeisconducting.
Assumingaforwardvoltagedrop,VF,of0.
6V,thepowerdissipatedinthediodecanbeapproximatedby(71).
(71)ThefinalcomponenttoconsiderisRR,whichisusedtoresetthecurrentsensetransformerduringtheoff-time.
SinceRCSismuchsmallerthanRR,thesecondaryRMScurrentalwaysflowstoRCSwhenthediodeisconducting.
Whenthecurrentsensediodeisnon-conducting,RRispresenttomaintaincurrentflowinginthetransformersecondarynecessaryforreset.
Thereforetheresetvolt-secondsaredeterminedbythevalueofRR.
RRshouldbeselectedsuchthatthetransformerresettimeisshorterthantheminimumresettimeofthepowertransformer,TPWR.
IncreasingRRhastheeffectofreducingtheresettimebutincreasingtheresetvoltage,causingadditionalvoltagestresstothecurrentsensingdiode.
Forminimalvoltagestressonthecurrentsensediode,anapproximationforRRisgivenby(72).
(72)Thetotalpowerdissipatedusingthecurrentsensetransformercannowbedeterminedby(73).
(73)(74)Comparingtheresultof(74)to(65),thepowerdissipatedusingthecurrentsensetransformertechniqueofFigure15resultsinonly134.
2mWoftotalpowerdissipationcomparedto1.
05WwhenacurrentsenseresistorisusedinserieswiththeQMAINMOSFETsource.
Thisisalmostalwaysthecaseforlowinputvoltage,highcurrentdesignapplicationsandevenforsomeofflineapplicationsitmaybeworthwhiletocomparethelossesforeachofthetwocurrentsensingtechniques.
PowerStageDesignwww.
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7SummaryofPowerStageLossesThepowerdissipatedbytheoutputinductorstillhastobecalculated.
Lookingatthedatasheetfortheinductorchosen,thewindingresistanceis2.
5milliohms.
(75)Thecorelossesaresosmallrelativelyspeakingthattheyareignored.
Thetotalfullloadpowerdissipation(froma100Wload)inonlythepowerstageissummarizedinFigure16andisestimatedtobeapproximately10.
75W,resultinginanestimatedfullloadefficiencyof90%.
ThepowerestimateofFigure16neglectsthelossesintheinputandoutputcapacitors,aswellasthelossintheQAUXMOSFET,buttheseareassumedtobeminimalwithinthescopeofthisestimation.
Figure16.
PowerStageLossEstimatewww.
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Thecurrentsenseinformationisderivedfromtheprimarysideasdiscussedintheprevioussection.
However,thedcerrorsignalnecessaryforthevoltageloopportionmustbefedbackfromthesecondarysidetotheprimaryside.
Crossingtheisolationboundarycanbeaccomplishedbyusingmagneticfeedbackoroptocouplerfeedback.
Sincetheoutputinductoralreadyprovidestheprimaryreferencedbootstrapbias,addingasecondcoupledwindingtogathertheerrorvoltagefeedbacksignalisnotdesirableforthisexample.
ThereforetokeepallthecomponentchoicesOTS,anoptocouplerisusedandisconfiguredasshowninFigure17.
Figure17.
OptocouplerFeedbackandSecondarySideCompensatorThereareseveralcomponentsthatareaddedtothefeedbacklooptoovercometheturnonovershootthatisinherenttoanintegratingfeedbackloop.
ThediscussionandanalysisneedtoselectandplacetheseisdiscussedinDefeatingTurnOnOvershootandwillnotbedealtwithhereotherthantoreferthereadertothearticlereference[14].
TheminimumvoltageonthefeedbackpintogetoutputpulsesisdefinedbytheCurrentSenseLevelShiftvoltagetimestheresistorratiowithinthedevicewhichyields(0.
48V*5)2.
4V.
TherecommendedusablevoltageattheFBpinoftheUCC2897Aiswithintherangeof2.
4VWhenVFBislessthen2.
4VtheUCC2897Aoperatesinapulseskippingmode.
Sincetheconverterhassynchronousrectificationtheconvertershouldnevergointopulseskippingmode.
Thismeansthatbecauseofthesyncrectifieractionthefeedbackpinshouldnevergobelow2.
4V.
Thereforeoverthefull2:1VINrange,theFBvoltagecanbeexpectedtochangeproportionallywithintherangeof2.
4V0V.
ThenextconsiderationisthatthereferencevoltageoftheUCC2897Acanonlysource5mAofcurrent.
SinceVREFisusedasthepull-upvoltagefortheoptocoupleroutput,themaximumallowableIREFoflessthan2mAduringoperation.
Arbitrarilywewillchoosea2kΩresistor.
(76)Fromequation(76)wearewellunderthedesignlimitduringoperationatminimumdutycycle.
OptocouplerVoltageFeedbackwww.
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com36SLUA535A–May2010–RevisedJan2020SubmitDocumentationFeedbackCopyright2010–2020,TexasInstrumentsIncorporatedUnderstandingandDesigninganActiveClampCurrentModeControlledConverterUsingtheUCC2897ATheSFH690BThasaCurrentTransferRatio(CTR)between100%and300%.
IftheoptocouplerisbiasedfortheminimumCTRof100%,thenthecurrent,IOPTO,shouldbeequaltotheresultof(77).
ThismeansthatitshouldoperateeveniftheCTRis100%.
(77)SincetheTLV431cansinkupto25mAofcathodecurrent,thereisplentyofheadroomfordrivingtheoptocoupler.
InordertominimizetheDCgainoftheoptocoupler,20percentofthemaximumTLV431currentisallowed.
Theoptocouplerbiasingresistor,ROPTO,canbedeterminedfrom(78).
VOPTOisselectedbasedupontheminimumtransformersecondaryvoltageof6Vminus1.
5Vofheadroomforasimpleseriespassregulatordesign.
SincetheprobabilityisthattheCTRwillbemoretothemiddlewewillassumeaCTRforcalculationof200%.
Allthiswilldoisreducetherequiredcurrentthroughthebiasingresistor.
Testinghoweverhighlightedanoiseimmunityproblemwiththeseriesregulatorundercertainloadconditionsandthecircuitwasmodifiedtoproduceavoltageofapproximately9voltsbeforetheseriesregulatorbutkeptthe4.
5VDClevelattheoutputoftheseriesregulator.
Thecurrentthroughtheoptocouplerphotodiodehastobeabletogofromamaximumof1.
3mAtonear0mA.
Theforwardvoltagedropoftheopto-couplerdiodeis1.
3Vat5mAandtheTLV431minimumusablevoltagewillbedefinedas1.
24V.
Theoreticallythereisapproximately2Vvariationneededonthesecondarysidetoachievea1.
3mAchangethroughtheoptocouplerandphotodiodebutbecausetheconditionsarenottheidealdefinedconditionsthiswascutto1/3.
(78)BasedontheselectedbiasingresistorsandtheminimumCTR,theminimumgainoftheoptocouplerisgivenby(79).
(79)SincethemaximumdutycyclerequiresthatthevoltageontheVfbpinapproachVrefthenthevoltageacrossthephotodiodemustapproachtheforwardvoltdropofthephotodiode.
Sincewehavesynchronousrectifierstheminimumcurrentthroughthecurrentsenseresistorontheprimarywillbehalftheripplecurrentontheoutputinductortranslatedtothecurrentsenseresistor.
ThevoltageonVfbwillbelessthantheVrefunderallnormaloperationalconditions.
Oncethecircuitisbuiltandtestedtheoverallcontrolloopneedstobeoptimized.
Sincethegainoftheoptocouplerispartoftheoverallconvertergain,theoptocouplerbiasingresistorsmaybeadjustedtooptimizethePWMfeedbackvoltage.
BecauseoftheundocumentedpoleoftheCTRathigherfrequencies(reference[15])provisionwasmadeforazerotobeaddedtotheoptocouplergainbyR30andC25.
Thesecomponentswillbeaddedifneededattest.
www.
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comCompensatingtheFeedbackLoop37SLUA535A–May2010–RevisedJan2020SubmitDocumentationFeedbackCopyright2010–2020,TexasInstrumentsIncorporatedUnderstandingandDesigninganActiveClampCurrentModeControlledConverterUsingtheUCC2897A6CompensatingtheFeedbackLoopTheoverallcontrolloopisshowninFigure18.
TheloopconsistsoffivegainblocksdenotedbyK,Gcl(s),Gf(s),Gc(s)andGopto(s).
Krepresentstheprimarysideoftheconverterandconsistsofthecurrentsensingcircuit,slopecompensationandfeedbackvoltageallusedascontrollinginputstothePWMcomparator.
TheUCC2897AincludesslopecompensationcircuitrythatisinternaltothecontroldevicebutexternallyprogrammablebyasingleresistorfromRSLOPEtogroundreference.
Adiscussionoftheeffectoftheactiveclamponthefeedbackloopisdiscussedinreference[8].
Figure18.
UCC2897AControlSchematicGcl(s)isthesecondorderresonanceeffectformedbetweenthetransformerprimarymagnetizinginductanceandclampcapacitor.
Thisgainhasa(1-D)2functionbecausetheonlytimethemagnetizinginductanceisconnectedtotheclampcapacitorisduringtheOFFtimeofthemainswitch.
Assuch,ithasnodirectimpactonthecontrolloopbutdoesinfluencethegainasit'seffectaffectstheresidualmagnetizingcurrentinthetransformerandwillimpactbothphaseandgainattheresonantfrequencyoftheprimarysidemagnetizinginductanceandtheclampcapacitor'scapacitance.
Thisresonantfrequencyisafunctionofthedutycycle.
Theauthorhasbeenunabletofindasatisfactorymathematicalmeansofaddingthistothecontrolloopequationhoweverthedocumentationclearlyindicatesthatthegaincrossovermustbebeforethisresonance.
Eventhoughthecontrolloopofavoltagemodeconverterusingthispowertopologydoesnotsufferfromthesamefeedbacklooplimitation,theimpactoftheresonanceoftheclampcapacitorandthemagnetizinginductanceonthevoltageacrosstheclampcapacitorandthepotentialforsaturationofthetransformerimposessimilarrestrictionsontheloopbandwidth.
Gf(s)isthesecondarysideofthepowerstageshownwiththeoutputinductorremoved.
Becausetheoutputinductorcurrentisoneofthecontrolvariables,thedoublepoleeffectnormallyseeninvoltagemodecontrolledconvertersisremovedthussimplifyingthecompensation.
Gc(s)isthesecondarysidecompensatorusingaTLV431setupinatype2configuration.
Becauseofitslowcost,theTLV431isaverypopularchoiceforuseastheerroramplifier.
CompensatingtheFeedbackLoopwww.
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com38SLUA535A–May2010–RevisedJan2020SubmitDocumentationFeedbackCopyright2010–2020,TexasInstrumentsIncorporatedUnderstandingandDesigninganActiveClampCurrentModeControlledConverterUsingtheUCC2897AGopto(s)istheoptocouplergainblockasdescribedintheprevioussection.
ThevaryingTLV431cathodevoltagesetsthediodecurrentoftheoptocoupler.
ThegainandCTRoftheoptocouplerdeterminetheemittercurrentseenontheprimaryside.
ThevaryingemittercurrentisthenusedtosettheDCcontrolvoltageseenbytheUCC2897A.
InsidetheUCC2897A,thefeedbackvoltageisbufferedanddivideddownby1/5beforetheinvertinginputofthePWMcomparator.
FromthecontrolschematicofFigure18,asimplifiedgainblockdiagramisshowninFigure19.
WiththeexceptionofGc(s),thecomponentsthatmakeupeachblockareknownandcannowbeusedtodefinethecontroltooutputtransfer,Gco(s).
Figure19.
UCC2897ASimplifiedControlBlockDiagramThegainconstantKissimplifiedanddefinedby(80).
TheadditionalvoltageplacedonthecurrentsensepinduetoslopecompensationalsohasasmallnegligibleeffectonK,butitisomittedhereforsimplification.
Theratiooftheinternalresistorswhichdividedownthecontrolsignaltomeetthevoltageofthecurrentsensepinisincludedhere.
Vcontrol(Vc)isthevoltageatthefeedbackpin.
(80)UniquetotheactiveclampoperatinginpeakCMC,isaresonanteffectoccurringbetweenthetransformermagnetizinginductanceandtheclampcapacitor.
Thiswillimpactthecontrolloopdesignandisdiscussedinmoredetailinreferences[7]and[8].
BecausetheclampcapacitorisonlyconnectedduringthetimethatQAUXisONitresultsinashiftintheresonanceasafunctionof1/(1-D)2.
Sincetheclampcapacitorisnottheonlycapacitancethatisinvolved,theimpactonthefrequencyisnotascleanastheequationsuggests.
Theeffectivecapacitanceofthesynchronousswitchesontheoutput,addintotheequationresultingintheresonantfrequencybeingloweredfurtherthantheanalysissuggests.
Wewillignorethemandrecognizethecauseofthetestedresultsoccurringatalowerthanexpectedfrequency.
Alsoignoredwillbeanycorelosseswhichwouldtendtodampenthepeakoftheresonance.
(81)(82)www.
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comCompensatingtheFeedbackLoop39SLUA535A–May2010–RevisedJan2020SubmitDocumentationFeedbackCopyright2010–2020,TexasInstrumentsIncorporatedUnderstandingandDesigninganActiveClampCurrentModeControlledConverterUsingtheUCC2897AThechangeoftheresonantfrequencyduetothedifferentdutycycleswithdifferentinputvoltagesareshowninFigure20.
Figure20.
EffectoftheMagnetizingInductanceandClampCapacitoronPartoftheControltoOutputGainLoopSincetheclampcapacitorisnotconnectedtothecurrentsenseresistoryoumightthinkthatthereisnoimpactfromthisresonancebutinrealityitimpactsthemagnetizingcurrentwhichformsapartofthecurrentthatissensed.
Toillustratethisimaginethattheconverterisinasteadystateminimumloadcondition.
Thishasthevoltageacrossthecapacitorinastableoscillation.
Nowassumethereisloadchangetomaximum.
ThecontrollooprespondsbyincreasingthedutycycleandthisresultsinanincreasedeltainthemagnetizingcurrentduringtheONtimeandlesstimeforadecreaseinthemagnetizingcurrentduringtheshortenedOFFtime.
Thisrepeatsoverseveralcyclecreatinganetincreaseinthemagnetizingcurrentandthevoltageacrosstheclampcapacitor.
Thechangedmagnetizingcurrentwhichbecomessignificantisaddedintothetotalcurrentbeingmonitored.
Thecontrolloopwhichhasincreasedtogetacertaincurrenttotheloadisdeliveringthatmagnitudeofcurrentbutadisproportionateamountofthatcurrentisthemagnetizingcurrent.
Thiseffectivelyreducesthecontroltooutputgainoftheloop.
Inadditionthephasedelayofthiscurrentwillalsoimpactthecontrolloopsphaseshift.
Bothoftheseeffectsarefeltattheresonantfrequencyoftheeffectiveclampcapacitorandthemagnetizinginductanceasafunctionof1/(1-D)2.
CompensatingtheFeedbackLoopwww.
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com40SLUA535A–May2010–RevisedJan2020SubmitDocumentationFeedbackCopyright2010–2020,TexasInstrumentsIncorporatedUnderstandingandDesigninganActiveClampCurrentModeControlledConverterUsingtheUCC2897ATherestofthecontrolloopisstraightforward.
Thetransferfunction,Gfoftheoutputfilterisreducedtoafirstordersystemgivenby(83).
ThisfunctionistheoutputimpedanceoftheloadinparallelwiththeoutputfilternormalizedabouttheDCimpedanceoftheoutputfilter.
Thereforeastheloadimpedancechangessodoesthegain.
InFigure21arethegainsasafunctionoffrequency,shownforthemaximumloadandforaloadthatis10%ofthemaximumDCload.
(83)Figure21.
GainoftheOutputLoadandCapacitorasaFunctionofFrequencyandRloadAsshowninFigure19,thecontroltooutputgainofthesystemisgivenby(84)andspecificallyinthefrequencydomainbyequation(85).
Theequationisafunctionoftheload(Rload)asshowninequation(85)(84)(85)Thecontroltooutputgainisplottedbelow.
Therearetwoseparateconditionsconsidered.
Theseareatfullloadandat10%load.
Theybreakdownasfollows:G1ismaximumload,G2is10%loadwww.
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comCompensatingtheFeedbackLoop41SLUA535A–May2010–RevisedJan2020SubmitDocumentationFeedbackCopyright2010–2020,TexasInstrumentsIncorporatedUnderstandingandDesigninganActiveClampCurrentModeControlledConverterUsingtheUCC2897ATheeffectsoftheresonanceoftheprimarysidemagnetizinginductancewiththeclampcapacitorareignoredasthereisnosatisfactorywaytomathematicallymodelthenbuttheywillbeafactorischoosingthe0dbcrossoverpointandtestresultswillbeusedtofinetunethecrossoverpoint.
Figure22.
GraphShowingthePlotoftheControltoOutputGainIncorporatingallthePreviousDefinedEquationsFrom(79),thedcgainoftheoptocoupler,Gopto,hasalreadybeencalculatedas12dB.
However,theoptocoupleralsoexhibitsanundocumentedsinglepolerolloffoccurringatroughly30kHz,andcanbecombinedwithGoptotogive(86),whichrepresentstheoptocouplerperformancewithfrequency.
(Seereference[15])Sincethesmallsignalresponseofanoptocouplerisnotspecifiedwithinthemanufacturer'sdatasheetandcanvaryforagivenapplication,itshouldbemeasuredincircuittovalidatetheassumptionsusedinthecontrolloopmodel.
(86)ForaforwardconverteroperatinginpeakCMC,atype2compensationnetworkisgenerallyused.
FortheCMCactiveclampforwardconverterthiscompensationschemecanbeusedwhentheoverallcrossoverfrequencyisdesignedtobeatleastafactorof5lowerthantheresonantfrequencyoftheclamp.
Thedesiredcrossoverfrequencyisdefinedby(88).
(87)(88)CompensatingtheFeedbackLoopwww.
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com42SLUA535A–May2010–RevisedJan2020SubmitDocumentationFeedbackCopyright2010–2020,TexasInstrumentsIncorporatedUnderstandingandDesigninganActiveClampCurrentModeControlledConverterUsingtheUCC2897AFrom(88)thecrossoverfrequency,F0,ofthecontrolloopisselectedtobe8kHz.
Byplotting(86)(optocouplergainassumingaCTRof200%)andaddingtheresulttoG1,thetotalclosedloopuncompensatedgainandphasearenowknownandthecompensationnetworkthatmakesupGc(s)inFigure18cannowbedesignedasfollows.
FromFigure23,andequation(89)theuncompensatedoverallgainisabout12.
5dBatF0=8kHz.
TherequiredabsolutegainatF0istheinverseofthatgivenby(89).
Thecompensatorneedstobedesignedtohavea–12.
5dBgainatthecrossoverfrequency.
(89)Figure23.
ControltoOutputgainFigure23showingtheControltoOutputgain,with(GdB)andwithout(G2dB),theopto-couplergainaddedin,andtheoptocouplergain(Gopto)allbyitself.
TheoptocouplergainisassumingaCTRof200%.
IfRX(Figure18)isarbitrarilychosentobe17.
4kΩ,thenRIcanbecalculatedfrom(90).
Actualcomponentvaluesusedinthefinaldesignareinthetextaftertheequation.
(90)RXistworesistorsinseriesR27is12.
1kandR28is4.
99kforatotalof17.
09kΩ.
TheinputresistorRIisalsomadeoftworesistorsR25whichis51.
1ΩandR26whichis28.
7kΩforatotalof28.
75kΩ.
Thisshouldresultinabout50mVmorethanthe3.
3Vandhelpcompensateconductionlossesintheconnector.
www.
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comCompensatingtheFeedbackLoop43SLUA535A–May2010–RevisedJan2020SubmitDocumentationFeedbackCopyright2010–2020,TexasInstrumentsIncorporatedUnderstandingandDesigninganActiveClampCurrentModeControlledConverterUsingtheUCC2897ATherequiredgainatcrossoveriscalculatedfrom(91)andthefeedbackresistor,RFB,ischosentoprovidetherequirednegativegainatF0andiscalculatedfrom(92)where.
(91)(92)TheactualvalueusedaftertestingforRFBwas5.
11ktodecreasethemeasuredloopgain.
ThemostprobablecauseofthehigherthancalculatedgainwastheoptocouplerCTR.
ThedesignassumedaCTRof200%andinrealityitisprobablysomewherebetween200%and300%.
Inaddition,azerowasaddedacrossR18toshiftthephaseatthecrossoverfrequencytoremovethephaseshiftingeffectofthepoleintheopto-coupler.
ThepoleformedbyRFBandCPisusedtorolloffthenoisefromtheswitchingfrequencyandissetathalftheoscillatorfrequencybytheformula(93).
ThisisstandardpracticeforaforwardCMCconverter.
(93)CPissetto220pF.
ThezeroformedbyRFBandCZisusedtoprovideadditionalphaseboostatF0.
Itwasdesirabletohavethephaseshiftfromtheerrorampessentiallyflatwhentheloopgoesthroughthe0dBpoint.
TothisendCZisdeterminedtoequalRFBat1/20ththecrossoverfrequencyFcofrom(94).
(94)Wechoose82nFforthevalue.
Thisgivesanequationforthegainoftheopampof(95)(95)Theopto-couplerhadazeroaddedbyaddingaseriesresistor/capacitoracrossRoptothatintroducedazeroatabout40kHzandthenapoleat400kHz.
Thisintroducedaphaseshiftthatboostedthephasestartingat4kHzandimprovedthegainmarginforbetterstabilityspecificallyoverthefrequencyofthedoublepoleperturbationsintroducedbytheactiveclampcapacitorandthemagnetizinginductance.
Equation96showsthegainequationfortheoptocouplerinthisconfigurationwithaCTRof100%whichistheminimum.
Ifweincreasethisto200%(assumednominal)or300%,thedefinedmaximum,wewillneedtomultiplythisby2or3respectively.
Thesegainoftheminimumopto-couplerfrequencyresponsewiththeadditionalzero/polecircuitisshowninfigures25A(gain)andB(phase).
CompensatingtheFeedbackLoopwww.
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com44SLUA535A–May2010–RevisedJan2020SubmitDocumentationFeedbackCopyright2010–2020,TexasInstrumentsIncorporatedUnderstandingandDesigninganActiveClampCurrentModeControlledConverterUsingtheUCC2897AFigure24showsthegainandphaseoftheopto-couplerandcomponents.
(96)Figure24.
GainandPhaseoftheOpto-CouplerandComponentswww.
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comCompensatingtheFeedbackLoop45SLUA535A–May2010–RevisedJan2020SubmitDocumentationFeedbackCopyright2010–2020,TexasInstrumentsIncorporatedUnderstandingandDesigninganActiveClampCurrentModeControlledConverterUsingtheUCC2897AThefinalcomponentvaluestestedintheactualdesignwereonlyslightlychangedfromcalculatedvaluesandareshownforcompletenessinFigure25.
Figure25.
Type2Compensator(FinalComponentDesignValuesShown)ThecalculatedgainandphaseresponsesofthecompensatedTLV431areshowninFigure26.
CompensatingtheFeedbackLoopwww.
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AlsoshowninFigure26,isabluelineindicatingthemaximumgainbandwidthproduct(GBW)oftheopenloopTLV431.
Forthisdesign,thecompensatednetworkiswellbelowtheGBWlimit,butitshouldbenotednonetheless.
Figure25showsthephasecompensatorboostofnearly0°atthedesignedcrossoverfrequency.
Figure26.
Type2CompensationGainandPhaseWiththecircuitofFigure24introducedintothedesign,andaddinginthegainandphaseshiftoftheoptocouplerthecalculatedtotalopenloopgainandphaseresponsesareshowninFigure27.
Thisdoesnotincludeanyperturbationcausedbythemagnetizinginductanceandclampcapacitorresonancebutdoesincludethenominalandextremesforthecurrenttransferratiosoftheoptocoupler.
FromtheloopgainresponseofFigure27,thecrossoverfrequencyof8kHzisachievedwithagainofabout40dBatlowfrequencygain.
Figure27.
CalculatedTotalOverallLoopGainandPhasewww.
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comProgrammingtheUCC2897APWMControlIC47SLUA535A–May2010–RevisedJan2020SubmitDocumentationFeedbackCopyright2010–2020,TexasInstrumentsIncorporatedUnderstandingandDesigninganActiveClampCurrentModeControlledConverterUsingtheUCC2897A7ProgrammingtheUCC2897APWMControlICUsingdesigninformationfromthepowerstage,thePWMcontrollercannowbesetup.
Thisisgenerallythefinalstepincompletingthepowerconverterdesign.
Thefollowingdesignequationsareintendedtocomplementthestepbystep,setupprocedureshownintheapplicationsectionofreferences[1]and[2].
Actualcomponentvaluesusedinthedesignareshowntotherightofeachresult.
Figure28.
UCC2897ASetUpDiagram7.
1Step1.
OscillatorFromthepowerstagedesignweknowthemaximumontimeandthedesiredfrequency.
Wewillarbitrarilysetthedelaytimeto0.
1s.
(97)(98)TheoscillatorfrequencyandmaximumdutycycleclamparesetbyRON,ROFF,andRdelayaccordingto(99)(100)and(101).
(99)(100)(101)Theactualnumbersusedwere8.
45KforRDEL,69.
8KforRONand88.
7KforROFF.
ProgrammingtheUCC2897APWMControlICwww.
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2Step2.
SoftStartThesoftstartcapacitorissetaccordingtoadesiredsoftstarttimeappliedto(102).
Forthisexampleasoftstarttimeof30msisarbitrarilychosen.
(102)Thistimeonlycoversthetimethatthecapacitorisgoingfrom2.
5voltsto4.
5volts.
InactualfacttheIcisactivebutnotswitchingfromthetimethethevoltageonthesoftstartcapacitorstartstoincreaseeventhoughnoswitchingispresent.
Thismeansthatthesoftstartcapacitorshouldbefullchargedabout60msaftertheVrefvoltagegoesto5volts.
Thesoftstopdischargecurrentisthesameasthechargecurrentsointhisgivesabout30msofsoftstoptime.
However,turn-onovershootprovedtobeaproblemandadditionalcircuitrywasaddedonthesecondarytoprovideanoverridetothissoftstartalthoughthesoftstartcapacitorwasleftinplaceasitisalsousedforsoftstopwhichpreventstheselfoscillationofthesynchronousrectifiersdescribedinreference[12].
Thesoftstartmethodisdiscussedindetailinreference[14].
7.
3Step3.
VDDBypassRequirementsFirstthehighfrequencyfiltercapacitoriscalculatedbasedongatechargeparametersofQMAINandQAUX.
Assumingthattheswitchingfrequencyrippleshouldbekeptbelow100mVacrossCHF,itsvaluecanbeapproximatedby(103).
FromthedatasheetfortheSI7846QG(QMAIN)is35nC,andfromtheIRF6216AUXMOSFETdatasheet,QG(AUX)isalso35nC.
(103)WhentheVDDreachestheminimumturnonvoltagetheVrefvoltagecomesupto5voltsandthedevicestartstodrawcurrenttopowertheinternalelectronics.
CurrentstartstoflowtotheCsscapacitorbuttheoutputswitchescannotstartuntilafterVssreaches2.
5volts.
AfterVssreaches2.
5voltstheoutputswitchescanstart.
ThedesignissuchthattheVsstakes30mstogofrom2.
5voltsto4.
5voltssoitwilltakeslightlylongertogofrom0Vto2.
5V.
Thisenergymustbeaddedtotheenergyneededtopowerthedeviceforthetimeittakestogofrom2.
5Vto4.
5V.
Thiswillbecalculatedin3equations.
Equation104willbetheenergyintheVrefcapacitor.
(104)Equation105willdealwiththeenergyfromthechargingoftheVrefcapacitortothestartofthemainswitches.
(105)www.
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Thenumberofswitchcyclesisthe30mstimestheoperatingfrequency.
Eachofthosecyclesbasedonthedeviceschoosenhastochargethegateinputcapacitorandithasbeencalculatedthatthisis2ampsofcurrentfor20nsandthatboththeQAUXandQMAINhavethesamerequirement.
(106)ThetotalchargeneededbythedevicetostarttheconverteristhesumofEquation104throughEquation106.
ThisgivesatotalchargecalculatedinEquation107andisequaltothechangeinchargerequiredtochangethecapacitorvoltage.
(107)Thischargehastobedeliveredbytheinputcapacitorduringachangeinvoltagefrom12.
2volts(minimumstartvoltage)and7.
8volts(12.
2voltslessminimumhysterisisvoltageof4.
4volts).
FromthisinformationthesizeofthecapacitorcanbedeterminedbyEquation108.
(108)ThedifferencebetweenEquation9andEquation108iscoveredbythedifferentsoftstarttimesandtheamountofchargeoneachofthegates.
AsstatedafterEquation9thiscapacitancedidnotincludethevariationsonIssandtheimpactthiswouldhaveonthesoftstarttime.
Ifwetakethatintoaccountthecapacitancerequirementincreasestoapproximately320F.
ThiswasfelttobetoolargeacapacitortoputonthetestboardasitwouldhavetakentoolongtochargesoaseriesregulatorfortheinputVccwasinstalled.
ProgrammingtheUCC2897APWMControlICwww.
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4Step4.
InputVoltageMonitoringTheamountofhysteresiscurrentfedbacktotheLINEUVcomparatorisfirstcalculatedbyEquation109.
(109)TheamountofhysteresisvoltageisspecifiedbythedifferencebetweenVONandVOFFinTable1,andisusedtocalculateRIN1fromEquation110.
(110)Thislowervalueofresistorwillinactualfactgiveusonlyabout0.
4voltsofhysteresis.
Whichiswhatweareseeingontheunit.
Thelow-sideresistoroftheLINEUVdividerisnoweasilycalculatedfromEquation111.
(111)7.
5Step5.
CurrentSenseFilteringandSlopeCompensationTheUCC2897APWMcontrollerusesaninternalslopecompensationschemethatisexternallyprogrammablebyappropriatelyselectingtworesistors,RFandRSLOPE.
Thecurrentsensefilterresistor,RF,isselectedbaseduponthechosencornerfrequencyofthelowpassfilterformedbyRFandCF.
Asastartingpoint,ageneralruleofthumbistoselectthecornerfrequencytobe10timestheswitchingfrequency.
Also,CFshouldbechosenbetweentherecommendedlimitsof47pF≤CF≤270pF.
ArbitrarilypickingCFequalto100pF,RFcanbedeterminedfromEquation112.
(112)TheactualvalueusedontheEVMis1.
82kΩ.
Thecloseststandardresistorvalueof536ΩischosenforRF.
Theoutputinductorcurrentslopemustnowbedefinedasitisreflectedfromthesecondary,backtotheprimaryandthentranslatedtoavoltageslopeseenacrossthecurrentsenseresistor,RCS.
Whenacurrentsensetransformerisused,thevoltageequivalentcompensationrampcanbedeterminedfromEquation113.
(113)(114)Forapplicationsthatdonotuseacurrentsensetransformer,Equation113canstillbeappliedbymakingtheNCStermequaltoone.
UsingthecalculatedvaluesofRFanddVL/dt,RSLOPEcannowbedeterminedfromEquation115.
FromthedatasheettheCtvoltagegoesfromzeroto2voltsinthetimethattheconvertermainswitchcanbeon.
(115)(116)Theactualvalueusedwas158kΩ.
www.
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Componentvaluesshownmaydifferslightlyfromcalculatedvalues.
AlsoshowninFigure29iseachmanufacturerandcomponentpartnumbercorrespondingtotheschematicshowninFigure29.
Figure29.
UCC2897ADesignExampleSchematicSchematicandListofMaterialswww.
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UCC2897ADesignExampleListofMaterialsCountRefDesValueDescriptionSizePartNumberMFR3C1,C2,C42.
2uFCapacitor,Ceramic,2.
2uF,100-V,X7R,20%1812C4532X7R2A225MTDK2C10,C1210uFCapacitor,Ceramic,16V,X5R,20%1206stdstd0C11openCapacitor,Ceramic,vvV,[temp],[tol]1210StdVishay1C132.
2uFCapacitor,Ceramic,25V,X5R,10%0805stdstd1C141000pFCapacitor,Ceramic,50V,X7R,20%0805stdstd1C1582nFCapacitor,Ceramic,50V,X7R,10%0805stdstd1C16270pFCapacitor,Ceramic,50V,X7R,10%0805stdstd3C18,C26,C271.
0uFCapacitor,Ceramic,25V,X7R,10%0805stdstd2C19,C20330uFCapacitor,POSCAP,9.
0mOhms,6.
3V,20%7343(D)6TPF330M9LSanyo0C22openCapacitor,Ceramic,50V,X7R,10%0805stdstd1C231.
5uFCapacitor,Ceramic,16V,X7R,10%0805stdstd1C258.
2nFCapacitor,Ceramic,50V,X7R,10%0805stdstd2C3,C170.
1uFCapacitor,Ceramic,50V,X7R,20%0805stdstd3C5,C21,C24100pFCapacitor,Ceramic,50V,NPO,10%0805stdstd2C6,C70.
22uFCapacitor,Ceramic,50V,X7R,20%0805stdstd1C810nFCapacitor,Ceramic,50V,X7R,20%0805stdstd1C933nFCapacitor,Ceramic,250V,X7R,10%1206stdstd4D1,D2,D3,D11BAT54Diode,Schottky,200-mA,30-VSOT23BAT54Vishay1D1013VDiode,Zener,13-V,150-mWSOT23BZX84C13-7-FDiodes1D4BAT54CDiode,DualSchottky,200-mA,30-VSOT23BAT54CVishay1D5BAS70-04LT1Diode,DualseriesSchottky,70-VSOT23BAS70-04LT1OnSemi1D65.
1VDiode,Zener,5.
1-V,350-mWSOT23BZX84C5V1Vishay2D7,D8BAS16Diode,Switching,200-mA,85-V,350-mWSOT23BAS16Fairchild1D9TLV431BAdjustableprecisionshuntregulator,0.
5%SOT23TLV431BQDBZTTI1J1VinTerminalBlock,2-pin,15-A,5.
1mm0.
40x0.
35ED500/2DSOST5J2,J3,J4,J5,J13131435300Adaptor,3.
5-mmprobeclip(or131-5031-00)3.
5-mm131-4353-00Tektronix6J6,J7,J9,J10,J11,J12TestPinPrintedCircuitPin,0.
043Hole,0.
3Length0.
0433103-1-00-15-00-00-08-0Mill-Max1J8VoTerminalBlock,4-pin,15-A,5.
1mm0.
80x0.
35ED500/4DSOST1L1PA0373Inductor,2-uH,1pri,1sec0.
920x0.
78PA0373Pulse1Q1IRF6216MOSFET,P-ch,150-V,2.
2-A,240-mOhmSO8IRF6216PBFIR1Q2Si7846DPMOSFET,N-ch,150-V,6.
7-A,50-milliohmS08Si7846DPVishay4Q3,Q4,Q5,Q7RJK0328DPBMOSFET,N-ch,30-V,60-A,1.
6-milliohmLFPAKRJK0328DPBRenesas1Q6MMBT2222ABipolar,NPN,40-V,600-mA,225-mWSOT23MMBT2222AVishay1Q8ZVN3320FMOSFET,N-ch,200-V,60-mA,25-ohmsSOT23ZVN3320FZetexwww.
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UCC2897ADesignExampleListofMaterials(continued)CountRefDesValueDescriptionSizePartNumberMFR1R18.
45KResistor,Chip,1/8W,1%0805stdstd1R1126.
7KResistor,Chip,1/8W,1%0805stdstd2R13,R172.
00KResistor,Chip,1/8W,1%0805stdstd4R14,R29,R31,R340Resistor,Chip,1/8W,1%0805StdStd1R18499Resistor,Chip,1/8W,1%0805stdstd1R19100KResistor,Chip,1/10W,1%0603stdstd1R269.
8KResistor,Chip,1/8W,1%0805stdstd1R21100KResistor,Chip,1/8W,1%0805stdstd1R225.
11KResistor,Chip,1/8W,1%0805stdStd2R24,R3310.
0KResistor,Chip,1/8W,1%0805stdstd2R25,R3051.
1Resistor,Chip,1/8W,1%0805stdstd1R2628.
7KResistor,Chip,1/8W,1%0805stdstd1R2712.
1KResistor,Chip,1/8W,1%0805stdstd1R284.
99KResistor,Chip,1/8W,1%0805stdstd1R388.
7KResistor,Chip,1/8W,1%0805stdstd1R32249KResistor,Chip,1/8W,1%0805stdstd0R35,R36openResistor,Chip,1/8W,1%0805stdstd1R379.
09KResistor,Chip,1/8W,1%0805stdstd6R4,R10,R15,R16,R20,R232.
21Resistor,Chip,1/8W,1%0805stdstd1R5158KResistor,Chip,1/8W,1%0805stdstd1R61.
82KResistor,Chip,1/8W,1%0805stdstd3R7,R8,R121.
00KResistor,Chip,1/8W,1%0805stdstd1R94.
64Resistor,Chip,1/8W,1%0805stdstd1T1P8208Transformer,CurrentSense,10-A,1:100SMDP8208Pulse1T2PA0810Transformer,HighFrequencyPlanarPlanarPA0810Pulse1U1UCC2897AIC,Current-ModeActiveClampPWMControllerPW20UCC2897APWTI1U2SFH690BTIC,Phototransistor,CTR100%-300%SOP4SFH690BTVishay1--PCB,3.
6Inx2.
7Inx0.
062InHPA348Any4--BumponRubberbumpontransparent,0.
44"x0.
2"0.
44"x0.
2"SJ53033MNotes:1.
TheseassembliesareESDsensitive,ESDprecautionsshallbeobserved.
2.
Theseassembliesmustbecleanandfreefromfluxandallcontaminants.
Useofnocleanfluxisnotacceptable.
3.
TheseassembliesmustcomplywithworkmanshipstandardsIPC-A-610Class2.
4.
Refdesignatorsmarkedwithanasterisk('**')cannotbesubstituted.
AllothercomponentscanbesubstitutedwithequivalentMFG'scomponents.
SuggestedDesignImprovementswww.
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ComponentreferencedesignationsinthefollowingnotesrefertotheschematicofFigure29.
9.
1OutputSyncRectifiersApossibledesignimprovementwouldbetomodifythedrivetotheoutputrectifyingFETSsothatifavoltageisappliedtotheoutputwhentheconverterisunpoweredtheoutputFETsdonotgointoselfoscillations.
9.
2OvercurrentShutdownAddacircuitthatwillshuttheconverterdownintheeventofanextendedovercurrentcondition.
Withthepresentmethodofpoweringtheprimarysidecontroltheconverterwillstayinpeakcurrentlimitandnotpowerdown.
9.
3ComponentChangesTheEVMfortheUCC2897Awasreleasedpriortothecompletionofthisexhaustiveanalysis.
AsaresulttherearesomechangestocomponentvaluesthatcouldbeincorporatedtomorecloselybringtheEVMinlinewiththespecification.
10ConclusionAstepbystepdesignprocedureofa3.
3V,100WactiveclampforwardconverteroperatinginpeakCMChasbeenshow.
ThedesignexampleisbaseduponusingtheUCC2897AActiveClampPWMCurrentModecontroller,howeverthepowerstagedesignprocedureisapplicabletoanylow-sideactiveclampforwardconverter.
TheconceptofZVShasbeenexplainedasitappliestotheactiveclampforwardtopology.
Thedetailsofthemajorcomponentlosseswithinthepowerstagehavealsobeenexamined.
www.
ti.
comReferences55SLUA535A–May2010–RevisedJan2020SubmitDocumentationFeedbackCopyright2010–2020,TexasInstrumentsIncorporatedUnderstandingandDesigninganActiveClampCurrentModeControlledConverterUsingtheUCC2897A11References1.
UCC289/1/2/3/4Current-ModeActiveClampPWMController,Datasheet(SLUS542)2.
UCC2897Current-ModeActiveClampPWMController,Datasheet(SLUS591A)3.
UCC3580/-1/-2/-3/-4SingleEndedActiveClampResetPWM,Datasheet,(SLUS292A)4.
SteveMappus,UCC2891EVM,48-Vto1.
3-V,30-AForwardConverterwithActiveClampReset,User'sGuidetoAccommodateUCC2891EVM,(SLUU178)5.
SteveMappus,ReferenceDesignPR265A48Vto3.
3VForwardConverterwithActiveClampResetUsingtheUCC2897ActiveClampCurrentModePWMController,(SLUU192)6.
48-VInput,3.
3V/100WattConverterwithUCC3580-1Controller,User'sGuidetoAccommodatePMP206_CReferenceDesign,(SLUU146)7.
A.
Fontán,S.
Ollero,E.
delaCruz,J.
Sebastián,PeakCurrentModeControlAppliedtotheForwardConverterwithActiveClamp,IEEE19988.
QiongLi,F.
C.
Lee,DesignConsiderationsoftheActiveClampForwardConverterwithCurrentModeControlduringLarge-SignalTransient,IEEE20009.
Q.
Li,F.
C.
LeeandM.
M.
Jovanovic,DesignConsiderationsofTransformerDCBiasofForwardConverterwithActiveClampReset,IEEEAPECProceedings,pp.
553-559,March14-19,199910.
JohnBottrill,MainSwitchActiveClampForwardConverterTransitionExamined,Bodo'sPowerSystems–April200911.
JohnBottrill,ADiscussionoftheActiveClampTopology,Bodo'sPowerSystems–September200712.
JohnBottrill,Turningoffofaconverterwithself-drivensynchronousrectifiersovershootBodo'sPowerSystems–May200613.
JohnBottrill,InputCapacitorConsiderationsEDNKorea–September200614.
JohnBottrill,DefeatingTurnOnOvershootPowerManagementDesignLine.
com–April2,200615.
JohnBottrill,OptocouplersandFeedbackLoopsofHighFrequencyPowerConvertersEDN–June19,2009RevisionHistorywww.
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com56SLUA535A–May2010–RevisedJan2020SubmitDocumentationFeedbackCopyright2010–2020,TexasInstrumentsIncorporatedRevisionHistoryRevisionHistoryNOTE:Pagenumbersforpreviousrevisionsmaydifferfrompagenumbersinthecurrentversion.
ChangesfromOriginal(*)toARevisionPageChangedToncalculation.
47ChangedRoffcalculation.
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